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Nov
SP6126
High-Voltage, Step Down Controller in TSOT6
LX GND FB
FEATURES Wide 4.5V - 29V Input Voltage Range Internal Compensation Built-in High Current PMOS Driver Adjustable Overcurrent Protection Internal soft-start 600kHz Constant Frequency Operation 0.6V Reference Voltage 1% output setpoint accuracy Lead Free, RoHS Compliant Package: Small 6-Pin TSOT
6
5
4
SP6126
6 PinTSOT
1
VIN
2
GATE
3
VDR
DESCRIPTION
The SP6126 is a PWM controlled step down (buck) voltage mode regulator with VIN feedforward and internal Type-II compensation. It operates from 4.5V to 29V making is suitable for 5V, 12V, and 24V applications. By using a PMOS driver, this device is capable of operating at 100% duty cycle. The high side driver is designed to drive the gate 5V below VIN. The programmable overcurrent protection is based on high-side MOSFET's ON resistance sensing and allows setting the overcurrent protection value up to 300mV threshold (measured from VIN-LX). The SP6126 is available in a space-saving 6-pin TSOT package making it the smallest controller available capable of operating from 24VDC supplies. TYPICAL APPLICATION CIRCUIT
VIN
Q1 Si2343DS 2 1 Vin Rs=1k LX C7 0.1uF 6 C4 22uF R1 200k, 1% 4 VFB GND 5 D1 1N4148 SHDN High=Of f
2008 Exar Corporation
C1 10uF
12V GND
Gate L1, IHLP-2525CZ 6.8uH, 60mOhm, 4.5A
VOUT 3.3V 0-2.0A
SP6126
3 VDR
Ds MBRA340T3G
RZ 2K CZ 62pF R2 44.2k, 1%
GND
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ABSOLUTE MAXIMUM RATINGS
These are stress ratings only, and functional operation of the device at these ratings or any other above those indicated in the operation sections of the specifications below is not implied. Exposure to absolute maximum rating conditions for extended periods of time may affect reliability. Input Voltage....................................................-0.3V to 30V Lx.............................................................-2V to 30V FB.................................................................-0.3V to 5.5V Storage Temperature.............................-65 C to 150 C Junction Temperature...................................-40C to 125C Lead Temperature (Soldering, 10 sec)...................300 C ESD Rating.................1kV LX, 2kV all other nodes, HBM
ELECTRICAL SPECIFICATIONS
Specifications are for TAMB=TJ=25C, and those denoted by apply over the full operating range, -40C< Tj <125C. Unless otherwise specified: VIN =4.5V to 29V, CIN = 4.7F.
PARAMETER UVLO Turn-On Threshold UVLO Turn-Off Threshold UVLO Hysterisis Operating Input Voltage Range Operating Input Voltage Range Operating VCC Current Reference Voltage Accuracy Reference Voltage Accuracy Reference Voltage Reference Voltage Switching Frequency Peak-to-peak ramp Modulator Minimum ON-Time Duration Minimum Duty Cycle Maximum Duty Cycle Gate Driver Turn-Off Resistance Gate Driver Pull-Down Resistance Gate Driver Pull-up Resistance VIN - VDR voltage difference Overcurrent Threshold LX pin Input Current OFF interval during hiccup Soft start time SHDN Threshold SHDN Threshold Hysteresis
MIN 4.2 4.0
TYP 4.35 4.2 0.2
MAX 4.5 4.4
UNITS V V V
CONDITIONS
0C< Tj <125C 0C< Tj <125C 0C< Tj <125C
4.5 7 0.3 0.5 0.5 0.6 0.588 510 0.6 600 VIN/5 40 100 50 4 3 4.5 270 25 3 0.9 300 30 100 5 1.0 100
29 29 3 2 0.612 690 100 0 60 8 6 5.5 330 35 9 1.1
V V mA % % V V kHz V ns % % k
VFB=1.2V

Internal resistor between GATE and VIN VIN=12V, VFB=0.5V, Measure resistance between GATE and VDR VIN=12V, VFB=0.7V, Measure resistance between GATE and VIN
V mV uA ms ms V mV
Measure VIN - VDR, VIN>7V Measure VIN - LX VLX = VIN VFB=0.58V, measure between VIN=4.5V and first GATE pulse
Apply voltage to FB
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PIN DESCRIPTION PIN # 1 2 3 PIN NAME VIN GATE VDR DESCRIPTION Input power supply for the controller. Place input decoupling capacitor as close as possible to this pin. Connect to the gate terminal of the external P-channel MOSFET. Power supply for the internal driver. This voltage is internally regulated to about 5V below VIN. Place a 0.1uF decoupling capacitor between VDR and VIN as close as possible to the IC. Regulator feedback input. Connect to a resistive voltage-divider network to set the output voltage. This pin can be also used for ON/OFF control. If this pin is pulled above 1V the P-channel driver is disabled and controller resets internal soft start circuit. Ground pin. This pin is used as a current limit input for the internal current limit comparator. Connect to the drain pin of the external MOSFET through an optional resistor. Internal threshold is pre-set to 300mV nominal and can be decreased by changing the external resistor based on the following formula: VTRSHLD = 300mV - 30uA * R BLOCK DIAGRAM
4 5
FB GND
6
LX
5V
VIN
VDR Oscillator Vin - 5V LDO
VIN 5V Internal LDO I = k x VIN
FAULT VREF
PWM Latch Reset Dominant S GATE
FB
+ + -
R
Error Amplifier
PWM Comparator VDR FAULT
FAULT ENBL UVLO FAULT Register S 4-Bit counter
-
LX
200ms delay
Overcurrent Comparator VIN - 0.3V
+ -
R R Set Dominant
POR
1V
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SP6126: TSOT-6 PFET Buck Controller
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+
30uA
GND
2008 Exar Corporation
GENERAL OVERVIEW The SP6126 is a fixed frequency, voltagemode, non-synchronous PWM controller optimized for minimum component, small form factor and cost effectiveness. It has been designed for single-supply operation ranging from 4.5V to 29V. SP6126 has Type-II internal compensation for use with Electrolytic/Tantalum output capacitors. For ceramic capacitors Type-III compensation can be implemented by simply adding an R and C between output and Feedback. A precision 0.6V reference, present on the positive terminal of the Error Amplifier, permits programming of the output voltage down to 0.6V via the FB pin. The output of the Error Amplifier is internally compared to a feed-forward (VIN/5 peak-topeak) ramp and generates the PWM control. Timing is governed by an internal oscillator that sets the PWM frequency at 600kHz. SP6126 contains useful protection features. Over-current protection is based on high-side MOSFET's Rds(on) and is programmable via a resistor placed at LX node. Under-Voltage Lock-Out (UVLO) ensures that the controller starts functioning only when sufficient voltage exists for powering IC's internal circuitry. SP6126 Loop Compensation The SP6126 includes Type-II internal compensation components for loop compensation. External compensation components are not required for systems with tantalum or aluminum electrolytic output capacitors with sufficiently high ESR. Use the condition below as a guideline to determine whether or not the internal compensation is sufficient for your design. Creating a Type-III compensation Network The above condition requires the ESR zero to be at a lower frequency than the double-pole from the LC filter. If this condition is not met, Type-III compensation should be used and can be accomplished by placing a series RC combination in parallel with R1 as shown below. The value of CZ can be calculated as follows and RZ selected from table 1.
CZ =
LC .............. (4) R1
fESRZERO / fDBPOLE
1X 2X 3X 5X >= 10X
RZ 50K 40K 30K 10K 2K
Table1- Selection of RZ
SP6126
Vout
CP1
2pF
RZ
Type-II internal compensation is sufficient if the following condition is met:
CZ CZ2 130pF RZ2 200k VFB R1 200k, 1%
f ESRZERO < f DBPOLE ................... (1)
where:
Error Amplif ier
f ESRZERO = f DBPOLE =
1 2. .R ESR .C OUT
........... (2) Figure 1- RZ and CZ in conjunction with internal compensation components form a Type-III compensation
1 2. . L C OUT
............ (3)
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SP6126: TSOT-6 PFET Buck Controller
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+
Vref =0.6V R2
2008 Exar Corporation
GENERAL OVERVIEW
Loop Compensation Example 1- A converter utilizing a SP6126 has a 6.8uH inductor and a 22uF/5m ceramic capacitor. Determine whether Type-III compensation is needed. From equation (2) fESRZERO = 1.45MHz. From equation (3) fDBPOLE = 13 kHz. Since the condition specified in (1) is not met, Type-III compensation has to be used by adding external components RZ and CZ. Using equation (4) CZ is calculated 61.2pF (use 62 pF). Following the guideline given in table 1, a 2k RZ should be used. The steps followed in example 1 were used to compensate the typical application circuit shown on page 1. Satisfactory frequency response of the circuit, seen in figure 2, validates the above procedure.
Loop Compensation Example 2- A converter utilizing a SP6126 has a 6.8uH inductor and a 150uF, 82m Aluminum Electrolytic capacitor. Determine whether Type-III compensation is needed. From equation (2) fESRZERO = 13kHz. From equation (3) fDBPOLE = 5 kHz. Since the condition specified in (1) is not met, Type-III compensation has to be used by adding external components RZ and CZ. Using equation (4) CZ is calculated 160pF (use 150 pF). Since fESRZERO / fDBPOLE is approximately 3, RZ has to be set at 30k .
Figure 2- Satisfactory frequency response of typical application circuit shown on page 1. Crossover frequency fc is 80kHz with a corresponding phase margin of 65 degrees.
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GENERAL OVERVIEW Overcurrent Protection Using the above equation there is good agreement between calculated and test results when an RS in the range of 0.5k to 3k is used. For RS larger than 3k test results are lower than those predicted by (5), due to circuit parasitics.
Vin
SP6126
Gate Q1
Using the ON/OFF Function Feedback pin serves a dual role of ON/OFF control. The MOSFET driver is disabled when a voltage greater than 1V is applied at FB pin. Maximum voltage rating of this pin is 5.5V. The controlling signal should be applied through a small signal diode as shown on page 1. Please note that an optional 10k bleeding resistor across the output helps keep the output capacitor discharged under no load condition. Programming the Output Voltage To program the output voltage, calculate R2 using the following equation:
Ov er-Current Comparator LX Rs
Figure 3- Overcurrent protection circuit
The overcurrent protection circuit functions by monitoring the voltage across the high-side FET Q1. When this voltage exceeds 0.3V, the overcurrent comparator triggers and the controller enters hiccup mode. For example if Q1 has Rds(on)=0.1 , then the overcurrent will trigger at I = 0.3V/0.1 =3A. To program a lower overcurrent use a resistor Rs as shown in figure 1. Calculate Rs from:
0.3 - (1.15 x Iout x Rds (on) ) Rs = ........... (5) 30uA
The overcurrent circuit triggers at peak current through Q1 which is usually about 15% higher than average output current. Hence the multiplier 1.15 is used in (5). Example: A switching MOSFET used with SP6126 has Rds(on) of 0.1 . Program the overcurrent circuit so that maximum output is 2A.
Rs =
0.3 - (1.15 x 1A x 0.1Ohm ) 30uA
Rs = 2333
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+ 30uA Vin - 0.3V
Ds
R2 =
R1 Vout Vref - 1
Where: Vref=0.6 is the reference voltage of the SP6126 R1=200k is a fixed-value resistor that, in addition to being a voltage divider, it is part of the compensation network. In order to simplify compensation calculations, R1 is fixed at 200k . Soft Start Soft Start is preset internally to 5ms (nominal). Internal Soft Start eliminates the need for the external capacitor CSS that is commonly used to program this function. MOSFET Gate Drive P-channel drive is derived through an internal regulator that generates VIN-5V. This pin (VDR) has to be connected to VIN with a 0.1uF decoupling capacitor. The gate drive circuit swings between VIN and VIN-5 and employs powerful drivers for efficient switching of the Pchannel MOSFET.
SP6126: TSOT-6 PFET Buck Controller
2008 Exar Corporation
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GENERAL OVERVIEW Power MOSFET Selection Select the Power MOSFET for Voltage rating BVDSS, On resistance RDS(ON), and thermal resistance Rthja. BVDSS should be about twice as high as VIN in order to guard against switching transients. Recommended MOSFET voltage rating for VIN of 5V, 12V and 24V is 12V, 30V and 40V respectively. RDS(ON) has to be selected such that when operating at peak current and junction temperature the Overcurrent threshold of the SP6126 is not exceeded. Allowing 50% for temperature coefficient of RDS(ON) and 15% for inductor current ripple, the following expression can be used: where:
Vf is diode forward voltage at IOUT Schottky's AC losses due to its switching capacitance are negligible. Inductor Selection Select the Inductor for inductance L and saturation current ISAT. Select an inductor with ISAT higher than the programmed overcurrent. Calculate inductance from:
300mV RDS (ON ) 1.5 x 1.15 x Iout
Within this constraint, selecting MOSFETs with lower RDS(ON) will reduce conduction losses at the expense of increased switching losses. As a rule of thumb select the highest RDS(ON) MOSFET that meets the above criteria. Switching losses can be assumed to roughly equal the conduction losses. A simplified expression for conduction losses is given by:
Vout 1 1 L = (Vin - Vout ) x x x Vin f Irip
where: VIN is converter input voltage VOUT is converter output voltage f is switching frequency IRIP is inductor peak-to-peak current ripple (nominally set to 30% of IOUT) Keep in mind that a higher IRIP results in a smaller inductor which has the advantages of small size, low DC equivalent resistance DCR, high saturation current ISAT and allows the use of a lower output capacitance to meet a given step load transient. A higher IRIP, however, increases the output voltage ripple and increases the current at which converter enters Discontinuous Conduction Mode. The output current at which converter enters DCM is 1/2 of IRIP. Note that a negative current step load that drives the converter into DCM will result in a large output voltage transient. Therefore the lowest current for a step load should be larger than 1/2 of IRIP. Output Capacitor Selection Select the output capacitor for voltage rating, capacitance and Equivalent Series Resistance (ESR). Nominally the voltage rating is selected to be twice as large as the output voltage. Select the capacitance to satisfy the specification for output voltage overshoot/undershoot caused by current step load. A steady-state output current IOUT 2 corresponds to inductor stored energy of 1/2 L IOUT .
Vout Pcond = Iout x RDS (ON ) x Vin
MOSFET's junction estimated from: temperature can be
T = (2 x Pc x Rthja ) + Tambient
Schottky Rectifier selection Select the Schottky for Voltage rating VR, Forward voltage Vf, and thermal resistance Rthja. Voltage rating should be selected using the same guidelines outlined for MOSFET voltage selection. For a low duty cycle application such as the circuit shown on first page, the Schottky is conducting most of the time and its conduction losses are the largest component of losses in the converter. Conduction losses can be estimated from:
Vout Pc = Vf x Iout x 1 - Vin
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GENERAL OVERVIEW A sudden decrease in IOUT forces the energy surplus in L to be absorbed by COUT. This causes an overshoot in output voltage that is corrected by power switch reduced duty cycle. Use the following equation to calculate COUT:
Input Capacitor Selection Select the input capacitor for Voltage, Capacitance, ripple current, ESR and ESL. Voltage rating is nominally selected to be twice the input voltage. The RMS value of input capacitor current, assuming a low inductor ripple current (IRIP), can be calculated from:
I 2 2 - I12 Cout = L x Vos 2 - Vout 2
Where: L is the output inductance I2 is the step load high current I1 is the step load low current Vos is output voltage including overshoot VOUT is steady state output voltage Output voltage undershoot calculation is more complicated. Test results for SP6126 buck circuits show that undershoot is approximately equal to overshoot. Therefore above equation provides a satisfactory method for calculating COUT. Select ESR such that output voltage ripple (VRIP) specification is met. There are two components to VRIP: First component arises from charge transferred to and from COUT during each cycle. The second component of VRIP is due to inductor ripple current flowing through output capacitor's ESR. It can be calculated from:
Icin = Iout x D(1 - D )
In general total input voltage ripple should be kept below 1.5% of VIN (not to exceed 180mV). Input voltage ripple has three components: ESR and ESL cause a step voltage drop upon turn on of the MOSFET. During on time capacitor discharges linearly as it supplies IOUT-Iin. The contribution to Input voltage ripple by each term can be calculated from:
V , Cin =
Iout x Vout x (Vin - Vout ) fs x Cin x Vin 2
V , ESR = ESR(Iout - 0.5Irip )
V , ESL = ESL
(Iout - 0.5Irip )
Trise
Where Trise is the rise time of current through capacitor Total input voltage ripple is sum of the above:
1 Vrip = Irip x ESR 2 + 8 x Cout x fs
Where: IRIP is inductor ripple current fs is switching frequency COUT is output capacitor calculated above
2
V , Tot = V , Cin + V , ESR + V , ESL
Note that a smaller inductor results in a higher IRIP, therefore requiring a larger COUT and/or lower ESR in order to meet VRIP.
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APPLICATION CIRCUITS
VIN
Q1 Si2343DS 2 1 Vin Rs=1k LX C7 0.1uF 6 L1, IHLP-2525CZ 6.8uH, 60mOhm, 4.5A Gate C1 10uF
12V GND VOUT
SP6126
3 VDR 4 VFB GND 5
Ds MBRA340T3G
RZ 2K R1 200k, 1% CZ 62pF R2 44.2k, 1%
C4 22uF
3.3V 0-2.0A
GND
D1 1N4148 SHDN High=Of f
Figure 4- Application circuit for VIN=12V
VIN
C1 2.2uF M1, Si4447DY 72mOhm, 40V 2 1 Vin Rs 1k LX C7 0.1uF 6 L1, Vishay IHLP-2525CZ 6.8uH, 4.5A, 60mOhm Gate C2 2.2uF
24-29V
GND
C1, C2 CERAMIC, 50V
VOUT
C4, ceramic 22uF, 6.3V R1 200k, 1%
SP6126
3 VDR 4 VFB GND 5
Ds, MBRA340T3 3A, 40V
RZ 2K CZ 62pF
3.3V 0-2.0A
GND
R2 44.2k, 1% SHDN D1 1N4148
Figure 5- Application circuit for VIN = 24-29V
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VIN
C1 10uF M1, Si2335DS 51mOhm, 12V 2 1 Vin Rs 1k LX C7 0.1uF 6 C4, ceramic 22uF, 6.3V R1 200k, 1% L1, Vishay IHLP-2525CZ 3.3uH, 6A, 30mOhm Gate C2 10uF C3 10uF
4.5-5.5 V
GND
VOUT 3.3V 0-3A
SP6126
3 VDR 4 VFB GND 5
Ds, MBRA340T3 3A, 40V
RZ 2K CZ 33pF
GND
R2 44.2k, 1% SHDN High=Of f D1 1N4148
Figure 6- Application circuit for VOUT = 4.5-5.5 V
TYPICAL PERFORMANCE CHARACTERISTICS
SP6126 Efficiency versus Iout, Vin=12V,Ta=25C
100
90 Efficiency (%)
80
70
Vout=3.3V Vout=5V Vout=2.5V
60 0.0 0.5 1.0 Iout (A) 1.5 2.0 2.5
Figure 7- Efficiency at VIN = 12 V
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TYPICAL PERFORMANCE CHARACTERISTICS
SP6126 Efficiency versus Iout, Vin=24V,Ta=25C
90
80 Efficiency (%)
70
Vout=3.3V
Vout=5V
60
50 0.0 0.5 1.0 Iout (A) 1.5 2.0 2.5
Figure 8- Efficiency at VIN = 24 V
SP6126 Efficiency versus Iout, Vin=5V,Ta=25C
100
Vout=2.5V Vout=3.3V
95
Efficiency (%)
90
85
80
75
70 0.0 0.5 1.0 1.5 Iout (A) 2.0 2.5 3.0
Figure 9- Efficiency at VIN = 5 V
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TYPICAL PERFORMANCE CHARACTERISTICS
Figure 10- Step load 1-2A, ch1: VIN; ch2: VOUT; ch3: IOUT
Figure 11- Step load 0.3-2A, ch1: VIN; ch2: VOUT; ch3: IOUT
Figure 12- Startup no load, ch1: VIN ch2: VOUT, ch3: IOUT
Figure 13- Start up 2A, ch1: VIN; ch2: VOUT; ch3: IOUT
Figure 14- Output ripple at 0A is 11mV, ch1: VIN; ch2: VOUT; ch3: IOUT
Figure 15- Output ripple at 2A is 18mV, ch1: VIN; ch2: VOUT; ch3: IOUT
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PACKAGE: 6PIN TSOT
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EXAR
SP6126: TSOT-6 PFET Buck Controller 2008 Exar Corporation
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ORDERING INFORMATION Part Number Temperature Range Package SP6126EK1-L........................................-40C to +125C...................(Lead Free) 6 Pin TSOT SP6126EK1-L/TR....................................-40C to +125C....................(Lead Free) 6 Pin TSOT
/TR = Tape and Reel Pack Quantity for Tape and Reel is 2500
For further assistance: Email: EXAR Technical Documentation: customersupport@exar.com http://www.exar.com/TechDoc/default.aspx? Exar Corporation Headquarters and Sales Office 48720 Kato Road Fremont, CA 94538 main: 510-668-7000 fax: 510-668-7030 EXAR Corporation reserves the right to make changes to the products contained in this publication in order to improve design, performance or reliability. EXAR Corporation assumes no responsibility for the use of any circuits described herein, conveys no license under any patent or other right, and makes no representation that the circuits are free of patent infringement. Charts and schedules contained here in are only for illustration purposes and may vary depending upon a user's specific application. While the information in this publication has been carefully checked; no responsibility, however, is assumed for inaccuracies. EXAR Corporation does not recommend the use of any of its products in life support applications where the failure or malfunction of the product can reasonably be expected to cause failure of the life support system or to significantly affect its safety or effectiveness. Products are not authorized for use in such applications unless EXAR Corporation receives, in writing, assurances to its satisfaction that: (a) the risk of injury or damage has been minimized; (b) the user assumes all such risks; (c) potential liability of EXAR Corporation is adequately protected under the circumstances.
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SP6126: TSOT-6 PFET Buck Controller
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